Controller for power convertor and motor driving system

ABSTRACT

A controller for a power convertor includes: a torque command value calculation module to calculate a first torque command value to a power convertor; a torque command limit module to receive the first torque command value and generate a second torque command value obtained by correcting the first torque command value so that the first torque command value is limited to a torque limiter range. The torque command limit module sets a width between the upper limit torque value and the lower limit torque value of the torque limiter range to be smaller as a fundamental wave output frequency of the power convertor increases at least in a speed region equal to or higher than a field weakening starting point.

FIELD

The present application relates to a controller for a variable voltageand variable frequency type power convertor and a motor driving system.

BACKGROUND

U.S. Pat. No. 9,281,772 discloses a controller for a power convertor.The controller can put, on a motor, a brake of good response. Thefollowing Non Patent Literature 1 discloses a flux observer for safeoperation of a motor.

Patent Literature 1: U.S. Pat. No. 9,281,772

Non Patent Literature 1: Y. Xu, Y. Wang, R. Iida and R. D. Lorenz,“Extending low speed self-sensing via flux tracking with volt-secondsensing,” 2017 IEEE Energy Conversion Congress and Exposition,Cincinnati, Ohio, 2017, pp. 1888-1895.

SUMMARY

However, in the systems of the above Patent Literature 1 and Non-PatentLiterature 1, no field weakening is implemented. For this reason, anoperation of the electric motor can be limited on the high speed range.

As a general technique of field weakening, there is a method ofsuppressing the influence of voltage saturation when the outputfrequency of the power converter increases, by lowering the flux commandvalue in inverse proportion to the rotational speed. The fundamentaloutput frequency of the power converter is also referred to as powersupply angular frequency.

The flux command method in simply inverse proportion to a frequency doesnot consider the output limit of the convertor. Therefore, the controlstability is potentially degraded when output torque in a fieldweakening region is increased.

The present application is intended to solve the above-described problemand provide a controller for a power convertor, and a motor drivingsystem that are capable of simultaneously achieving control stabilityand output torque increase in a field weakening region.

In the control device described in Non-Patent Literature 1, sensorlesscontrol is performed by obtaining the rotation speed using the statorflux estimated by the flux observer. However, this stator flux issubject to harmonics caused by PWM control and current measurementnoise. For this reason, the stator flux has a low S/N ratio because theflux amplitude decreases particularly in the field weakening region. Forthese reasons, the accuracy of the speed estimation declines and thereis a possibility that stable operation cannot be performed.

The present application is also intended to provide a controller for apower convertor, and a motor driving system that are capable of highlyaccurately calculating an estimated speed value.

A first controller for a power convertor according to a first aspect ofthe present application includes: a torque command value calculationmodule configured to calculate a first torque command value to a powerconvertor based on a speed command value of a motor driven by the powerconvertor; a torque command limit module configured to receive the firsttorque command value and generate a second torque command value obtainedby correcting the first torque command value so that the first torquecommand value is limited to a torque limiter range defined by an upperlimit torque value calculated by a predetermined calculation formula forupper limit torque command value calculation and a lower limit torquevalue obtained by multiplying the upper limit torque value by apredetermined negative coefficient or zero; a flux command generationmodule configured to generate a stator flux command value in accordancewith a fundamental wave output frequency of output from the powerconvertor; and an output voltage calculation module configured tocalculate an output voltage command value of the power convertor basedon the second torque command value and the stator flux command value.The torque command limit module calculates the upper limit torque valueto be smaller as the fundamental wave output frequency increases atleast in a speed region equal to or higher than a field weakeningstarting point.

A second controller for a power convertor according to a second aspectof the present application includes: a torque command value calculationmodule configured to calculate a torque command value to a powerconvertor based on a speed command value of a motor driven by the powerconvertor; a voltage command value calculation module configured tocalculate a voltage command value to the power convertor based on thetorque command value calculated by the torque command value calculationmodule; a flux estimation module configured to calculate estimatedvalues of the stator flux and rotor flux of the motor in a subsequentcontrol period based on the voltage command value to the power convertorand a measured stator current of the motor; and a motor speed estimationmodule configured to calculate an estimated value of a speed of themotor in a subsequent control period based on the estimated value of therotor flux calculated by the flux estimation module.

A motor driving system according to a third aspect of the presentapplication includes: a power convertor configured to drive a motor; andone of the first or second controller for a power convertor, which isconfigured to control the power convertor.

Other and further objects, features and advantages of the invention willappear more fully from the following description.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a configuration diagram of a motor system to which acontroller for a power convertor according to a first embodiment isapplied;

FIG. 2 is an enlarged configuration diagram of part of the controllerfor a power convertor according to the first embodiment;

FIG. 3 is an enlarged configuration diagram of part of the controllerfor a power convertor according to the first embodiment;

FIG. 4 is a graph illustrating a stator flux locus for each power supply(operation) frequency with a voltage limit taken into account;

FIG. 5 is a graph illustrating a stator flux locus with a current limittaken into account;

FIG. 6 is a graph illustrating stator flux loci with the voltage andcurrent limits taken into account;

FIG. 7 is a graph illustrating voltage and current limits and torquecurves at a power supply frequency of 1.2 pu;

FIG. 8 is a graph illustrating maximum torque curves with the voltageand current limits taken into account at different speeds in Region I;

FIG. 9 is a graph indicating a boundary speed condition between Region Iand Region II;

FIG. 10 is a graph illustrating maximum torque curves with the voltageand current limits taken into account at different speeds in Region II;

FIG. 11 is a configuration diagram of a motor system to which acontroller for a power convertor according to a second embodiment isapplied;

FIG. 12 is an enlarged configuration diagram of part of the controllerfor a power convertor according to the second embodiment;

FIG. 13 is a diagram showing an example of a result of field weakeningtest in the second embodiment;

FIG. 14 is a diagram showing an example of a result of field weakeningtest in the second embodiment;

FIG. 15 is a diagram showing an example of a result of field weakeningtest in the second embodiment;

FIG. 16 is a configuration diagram of a motor system to which acontroller for a power convertor according to a modification of thesecond embodiment is applied;

FIG. 17 is an enlarged configuration diagram of part of the controllerfor a power convertor according to the modification of the secondembodiment;

FIG. 18 is a configuration diagram of a motor system to which acontroller for a power convertor according to a third embodiment isapplied;

FIG. 19 is a configuration diagram of a motor system to which acontroller for a power convertor according to a modification of thethird embodiment is applied;

FIG. 20 is a diagram illustrating a hardware configuration applicable tothe controller for a power convertor according to the first embodiment;and

FIG. 21 is a graph of relative RMS noise as flux decreases.

DESCRIPTION OF EMBODIMENTS

Embodiments of the present invention will be described in accordancewith the accompanying drawings. It should be noted that in the drawings,the same or corresponding parts are denoted by the same reference signs.Overlapping description of such parts will be simplified or omitted asappropriate.

Symbols are explained below. Complex vectors are as follows.

V_(qds) stator voltage complex vector [V]

V_(qdr) rotor voltage complex vector [V]

i_(qds) stator current complex vector [A]

i_(qdr) rotor current complex vector) [A]

λ_(qds) stator flux linkage complex vector [V-sec]

λ_(qdr) rotor flux linkage complex vector [V-sec]

Scalars are as follows.

V_(qs) stator q-axis voltage [V]

V_(ds) stator d-axis voltage [V]

V_(qr) rotor q-axis voltage [V]

V_(dr) rotor d-axis voltage [V]

i_(qs) stator q-axis current [A]

i_(ds) stator d-axis current [A]

i_(qr) rotor q-axis current [A]

i_(dr) rotor d-axis current [A]

λ_(qs) stator q-axis Flux [V-sec]

λ_(ds) stator d-axis Flux [V-sec]

λ_(qr) rotor q-axis Flux [V-sec]

λ_(dr) rotor d-axis Flux [V-sec]

T_(e) electromagnetic torque [N-m]

T_(L) load torque [N-m]

v_(us) u-phase stator voltage [V]

v_(vs) v-phase stator voltage [V]

v_(ws) w-phase stator voltage [V]

i_(us) u-phase stator current [A]

i_(vs) v-phase stator current [A]

i_(ws) w-phase stator current [A]

i_(ur) u-phase rotor current [A]

i_(vr) v-phase rotor current [A]

i_(wr) w-phase rotor current [A]

R_(s) stator resistance [Ω]

R_(r) rotor resistance [Ω]

L_(ls) stator leakage inductance [H]

L_(lr) rotor leakage inductance [H]

L_(m) magnetizing inductance [H]

L_(s) stator winding inductance [H] L_(s)=L_(ls)+L_(m)

L_(r) rotor winding inductance [H] L_(r)=L_(lr)+L_(m)

σ   leakage  factor    σ := 1 − (L_(m)²/L_(s)L_(r))

J_(p) inertia [kg-m²]

P Pole number

P/2 Pole pair number

t_(s) sampling time [s]

P_(cu) IM cupper loss [W]

P_(fe) IM iron loss [W]

P_(loss) IM loss [W] P_(loss)=P_(cu)+P_(fe)

P_(e) input power [W] P_(e)=P_(loss)+P_(stored)+P_(em)

P_(stored) Time differentiation of magnetic energy accumulated in theinductance of the induction machine [W]

P_(em) mechanical loss [W] (Power contributing to torque)

K_(e) eddy current coefficient

K_(h) hysteresis coefficient

R_(eq) equivalent resistance [Ω]

τ_(eq) equivalent time constant [s]

τ_(r) rotor time constant [s] That is, MI_T2 (Secondary time constant)

ω angular speed [rad/s] ω=pθ

General Description without Specifying Coordinate System (ArbitraryCoordinate System)

θangle [rad] ω=pθ General description without specifying the coordinatesystem

ω_(r) rotor electrical angular speed [rad/s] ω_(r)=Pω_(rm)/2

θ_(r) rotor electrical angular position [rad]

(Angle of u-Phase Rotor Winding Taken Counterclockwise with Respect tou-Phase Stator Winding)

ω_(rm) rotor mechanical angular speed [rad/s] ω_(rm)=2ω_(r)/P

Rotational Angular Speed of the Output Shaft

θ_(rm) rotor mechanical angle [rad] θ_(rm)=2θ_(r)/P

Rotational Angular of the Output Shaft

ω_(e) synchronous angular frequency [rad/s]

ω=pθ, but if the synchronization angular frequency is constant, it isalso expressed as ωt=θ.

θ_(e) synchronous angular position

ω_(sl) slip angular frequency [rad/s]

ω_(sl)=ω_(e)−ω_(r)

Mathematical Elements are as Follows.

j imaginary number

p differential operator

s Laplace operator

a space vector rotation operator a=exp(j2π/3)

Re{ } complex real part

Im{ } complex imaginary part

L{ } Laplace transform

Z{ } z-transform)

Each symbol of superscript has the following meaning.

f^(s) stationary reference frame

f^(e) general synchronous reference frame

f^(r) rotor reference frame

f^(ras) re-aligned stationary reference frame

f* command value

{dot over (f)} differential value

{circumflex over (f)} estimate value

f average value

Each subscript symbol has the following meaning.

f_(s) stator

f_(r) rotor

( )_(opt) appropriate value

First Embodiment

FIG. 1 is a configuration diagram of a motor system to which acontroller for a power converter according to an embodiment 1 of thepresent invention is applied.

In FIG. 1, a motor system 1 includes a motor 2 and a motor drivingsystem 3. For example, the electric motor 2 is an induction machine.

An output part of the motor 2 is connected to an input part of a loadmachine 4. For example, the load machine 4 is an inertia load. A speedsensor 119 for detecting the rotational speed of the rotor is connectedto the electric motor 2.

An input part of the motor driving system 3 is connected to an outputpart of an AC power supply 5. For example, the AC power supply 5 is agrid.

The motor driving system 3 includes a diode rectifier 6, a capacitor 7,an inverter 8, a first current detector 9 a, a second current detector 9b, and controller 11.

The diode rectifier 6 converts, into DC power, the three-phase AC powersupplied from the AC power supply 5. If necessary, the diode rectifier 6may be replaced with a PWM converter.

The capacitor 7 is provided across a DC link on the output side of thediode rectifier 6. The capacitor 7 is used to smooth the DC voltageapplied to the DC link.

The inverter 8 is converts the DC power supplied from the dioderectifier 6 into three-phase AC power for driving the electric motor 2.The inverter 8 is a voltage source inverter. The inverter 8 is subjectedto variable voltage variable frequency (VVVF) control through pulsewidth modulation (PWM) control.

The power conversion circuit of the inverter 8 is formed of three arms.One of the arms includes an upper arm and a lower arm. The upper andlower arms are each formed of at least one switching element.

The first current detector 9 a is provided at the v-phase of the outputside of the inverter 8. The first current detector 9 a detects thev-phase stator current Ivs. The second current detector 9 b is providedat the w-phase of the output side of the inverter 8. The second currentdetector 9 b detects the w-phase stator current Iws.

The controller 11 includes a speed controller 12, a DB-DTFC calculationmodule 14, a first coordinate conversion module 15, PWM controller 16, asecond coordinate conversion module 17, a current/flux estimation module20, a speed/phase estimation module 21, a torque command limit module13, an appropriate flux command generation module 18, a power supplyangular frequency calculation module 19, and a first slip angularfrequency estimation module 32.

The speed controller 12 is a torque command value calculation module.The speed controller 12 calculates the torque command value T_(em1)* sothat the rotor mechanical angular speed estimated value {circumflex over(ω)}_(rm-r), which is detected by the speed/phase estimation module 21,will follow the rotor mechanical angular speed command value ω_(rm)*obtained from an external device.

Here, the “dqs-axes” is such that the U phase and the q axis coincide ofa stator with each other in a stationary axis system, and distributesthe three phase components to two phases of the d axis and the q axisorthogonal to each other. The second coordinate conversion module 17,which will be described later, is a circuit that performs dq-axesconversion to replace the input signal with the biaxial signal. Thefirst coordinate conversion module 15, which will be described later, isan inverse conversion circuit that restores the two phase signalsconverted into the d axis and the q axis in the stationary axis systemto three phase signals in the stationary coordinate system.

For example, a superscript symbol “S” of the stator dqs-axes flux valueλ_(qds) ^(S) represents the stationary coordinate system. A subscriptsymbol “qds” of the stator dqs-axes flux value λ_(qds) ^(S) represents“two phase components” of the stator flux. That is, the stator dqs-axesflux value λ_(qds) ^(S) represents the d-axis component λ_(ds) ^(S) ofthe stator flux value and the q-axis component λ_(qs) ^(S) of the statorflux command value. In the following description, symbols with thesubscript suffix “qds” are the symbols representing both the d-axiscomponent and the q-axis component of the stator flux.

On the other hand, a superscript symbol “S” of the rotor dqs-axes fluxvalue λ_(qdr) ^(S) represents the stationary coordinate system. Asubscript symbol “qdr” of the rotor dqs-axes flux value Δ_(qdr) ^(S)represents the d-axis component λ_(dr) ^(S) of the rotor flux value andthe q-axis component λ_(qr) ^(S) of the rotor flux value. In thefollowing description, symbols with the subscript suffix “qdr” are thesymbols representing both the d-axis component and the q-axis componentof the rotor flux.

The torque command limit module 13 calculates the second torque commandvalue T_(em)* based on the first torque command value T_(em1)*calculated by the speed control module 12, the stator flux command valueλ_(s_opt) generated by the appropriate flux command generation module18, and the power supply angular frequency ω_(e) calculated by the powersupply angular frequency calculation module 19. Details of the torquecommand limit module 13 will be described later with reference to FIG.2.

The DB-DTFC calculation module 14 calculates the stator dqs-axes voltagecommand value V_(qds) ^(S)* based on the second torque command valueT_(em)*, the flux command value λ_(s)*, the stator dqs-axes fluxestimated value {circumflex over (λ)}_(ds) ^(S) the rotor dqs-axes fluxestimated value {circumflex over (λ)}_(qdr) ^(S), and the rotormechanical angular speed estimated value {circumflex over (ω)}_(rm-r)estimated by the speed/phase estimation module 21 which is describedlater. The flux command value λ_(s)* is a stator flux command valueλ_(s_opt) generated by the appropriate flux command generation module18. The DB-DTFC calculation module 14 employs, as a control method, adeadbeat direct torque & flux control (DB-DTFC) method.

The first coordinate conversion module 15 converts the stator dqs-axesvoltage command value V_(qds) ^(S)* into three-phase stator voltagecommand values V_(us)*, V_(vs)*, and V_(ws)*. The conversion performedby the first coordinate conversion module 15 is an inverse conversion ofthe dqs-axes transformation.

The PWM controller 16 converts the three-phase stator voltage commandvalues V_(us)*, V_(vs)*, and V_(ws)* into gate pulses for the inverter 8based on the pulse width modulation. The PWM controller 16 outputs thegate pulses to the inverter 8.

The second coordinate conversion module 17 converts the stator currentsIvs, Iws into a stator dqs-axes current measured value i_(qds) ^(s). Theconversion performed by the second coordinate conversion module 17 is“dqs-axes conversion”.

Three-phase to two-phase conversion by dqs-axes conversion is performedbased on the following equation. For example, if the currents of threephases are I_(u), I_(v), I_(w) and the currents after two-phaseconversion are I_(ds), I_(qs), the following equation is obtained.I _(u) +I _(v) +I _(w)=0I _(qs) =I _(u)I _(ds)=(I _(u)+2I _(w))/√3

This conversion is somewhat different from Clarke conversion which is ageneral three-phase to two-phase conversion.

The appropriate flux command generation module 18 generates the statorflux command value λ_(s_opt) by performing calculation on an appropriateflux command based on the power supply angular frequency ω_(e)calculated by the power supply angular frequency calculation module 19.The stator flux command value λ_(s_opt) having an appropriate value isinput to the DB-DTFC calculation module 14 as the flux command valueλ_(s)*. The appropriate flux command generation module 18 will bedescribed later in detail with reference to FIG. 2.

The power supply angular frequency calculation module 19 calculates thepower supply angular frequency ω_(e) based on the rotor mechanical angleestimated value {circumflex over (ω)}_(rm-r) estimated by thespeed/phase estimation module 21 and the slip angular frequencyestimated value {circumflex over (ω)}_(sl) estimated by the first slipangular frequency estimation module 32. The relationship between thedetected rotational speed ω _(rm) from the speed sensor 119 and theestimated value ω_(rm-r) will be described later with reference to FIG.3. Details of the power supply angular frequency calculation module 19will be described later with reference to FIG. 2.

It should be noted that the output value of the speed sensor 119 is notthe instantaneous value but the average speed between the detectiontimings, and thus the output value of the speed sensor 119 is expressedas ω _(rm) using the symbol indicating the average value.

Here, the stator dqs-axes current measured value i_(qds) ^(s) and thestator dqs-axes voltage measured value V_(qds) ^(s) are values obtainedby dqs converting the three phase measured values. The total sum of theinstantaneous values of the three-phase currents, which is the output ofthe inverter 8, is zero. By using this property, in the firstembodiment, two-phase current is detected, and for the other phase, avalue obtained by inverting the sign of the added value of two-phasecurrent is used.

The current/flux estimation module 20 calculates the stator dqs-axesflux estimated value {circumflex over (λ)}_(qds) ^(S) and the rotordqs-axes flux estimated value {circumflex over (λ)}_(qdr) ^(S) as a fluxestimation module based on the rotor electrical angle estimated value{circumflex over (θ)}_(r) estimated by the speed/phase estimation module21, the voltage command value V_(qds) ^(S)*, the stator dqs-axes currentmeasured value i_(qds) ^(s), the rotor mechanical angular speedestimated value {circumflex over (ω)}_(rm-r).

A speed and phase estimation module 21 as a motor speed estimationmodule calculates a rotor electrical angle estimated value {circumflexover (θ)}_(r) and a rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r) based on the second torque command valueT_(em)* and a measurement value ω _(rm) obtained by a speed sensor 119.The speed and phase estimation module 21 is what is called a motionobserver, and the configuration thereof is illustrated in FIG. 3. Themotion observer estimates a speed and a phase at subsequent samplingbased on a speed of the motor 2 or detected phase obtained from anoutput from the speed sensor 119 and a torque command. The motionobserver can obtain an instantaneous speed of the motor 2 at eachsampling without phase delay.

The speed/phase estimation module 21 basically performs a calculationsuch that a torque command value is divided by a moment of inertia ofthe motor 2 to calculate an acceleration value, the acceleration valueis integrated to calculate a speed, and the speed is integrated tocalculate a phase. The delay operator is used in the speed/phaseestimation module 21.

The rotational speed ω _(rm) of the motor 2, which is an output of thespeed sensor 119, is inputted into the subtraction module 21 a. Thesubtraction module 21 a calculates a difference between the rotationalspeed ω _(rm) and an output of the delay calculation module 21 m. Thecalculated difference is transmitted to the integral module 21 b and theproportional module 21 e. The integral module 21 b integrates an outputof the subtraction module 21 a. The integrated value by the integralmodule 21 b is transmitted to the integral module 21 c and theproportional module 21 d.

The integral module 21 c is an integration circuit having a gain Kio.The integral module 21 c outputs a calculation result to the addermodule 21 f. The proportional module 21 d performs amplification by again K_(so). The proportional module 21 d outputs a calculation resultto the adder module 21 f. The proportional module 21 e performsamplification by a gain bo. The proportional module 21 e outputs acalculation result to the adder module 21 f.

The adder module 21 f performs addition of the output from the integralmodule 21 c, the output from the proportional module 21 d, and theoutput from the proportional module 21 e, and outputs the calculationresult to the adder module 21 g. The adder module 21 g performs additionof the second torque command value T_(em)*, which is the output from thetorque command limit module 13, and the output from the adder module 21f, and then outputs a calculation result to the acceleration valuecalculation module 21 h.

The acceleration value calculation module 21 h divides an inputted valueby the moment of inertia estimated value ĵ_(p) of the motor 2, and thenoutputs the estimated acceleration value to the integral module 21 j.The integral module 21 j integrates the inputted estimated accelerationvalue to estimate the rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r).

The rotor mechanical angular speed estimated value {circumflex over(ω)}_(rm-r) is transmitted to the integral module 21 n and the averagevalue calculation module 21 k. The rotor mechanical angular speedestimated value {circumflex over (ω)}_(rm-r) is also transmitted to theoutside of the speed/phase estimation module 21. The outside of thespeed/phase estimation module 21 is for example the DB-DTFC calculationmodule 14. Further, the rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r) is also transmitted to the current and fluxestimation module 20, the power supply angular frequency calculationmodule 19, and the speed control module 12.

The integral module 21 n integrates the rotor mechanical angular speedestimated value {circumflex over (ω)}_(rm-r) to calculate the estimatedphase (mechanical angle) {circumflex over (θ)}_(rm). The estimated phase(mechanical angle) {circumflex over (θ)}_(rm) is inputted to theproportional module 21 p. The proportional module 21 p multiples theestimated phase (mechanical angle) {circumflex over (θ)}_(m) by the polenumber P/2 of the motor 2 to calculate the estimated phase (electricalangle) {circumflex over (θ)}_(r). The estimated phase (electrical angle){circumflex over (θ)}_(r) is outputted to the outside of the speed/phaseestimation module 21. The outside of the speed/phase estimation module21 is for example the current and flux estimation module 20.

The average value calculation module 21 k calculates the average value{circumflex over (ω)}_(rm) with respect to the rotor mechanical angularspeed estimated value {circumflex over (ω)}_(rm-r) by calculating anaverage of an estimated value in the present calculation period andanother estimated value before one calculation period, and then outputsthe calculation result to the delay calculation module 21 m. The delaycalculation module 21 m performs a calculation to delay the averagevalue {circumflex over (ω)}_(rm) by one calculation period, and thenoutputs a calculation result to the subtraction module 21 a as asubtraction value. It should be noted that, the configuration in FIG. 3may be modified in other system, and the output from the speed sensor119 may be a low pass filter, and an output from the low pass filter maybe regarded as the rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r).

Voltage, current, and flux always change in the dqs-axes coordinate inthe steady state. On the other hand, speed is constant, and theinfluence of the phase delay need not to be considered. By using themotion observer, phase and speed information with little noise can beobtained without phase delay.

FIG. 2 is an enlarged configuration diagram of part of the controllerfor a power convertor according to the first embodiment. As illustratedin FIG. 2, this controller 11 includes the speed control module 12, thetorque command limit module 13, the appropriate flux command generationmodule 18, and the power supply angular frequency calculation module 19.

The torque command limit module 13 includes a first block 13 a, a secondblock 13 b, a third block 13 c, a fourth block 13 d, a fifth block 13 e,and a sixth block 13 f.

The first block 13 a calculates an upper limit torque command valueT_(e_max) based on the stator flux command value λ_(s_opt) in accordancewith Expression (30) to be described later. Expression (30) is apredetermined first calculation formula for upper limit torque commandvalue calculation. A second-order polynomial including the second-orderterm (λ_(s_opt) ²) of the stator flux command value λ_(s_opt) and thefirst-order term (λ_(s_opt)) thereof is derived by expanding theexpression in the square root in Expression (30).

The second block 13 b calculates the upper limit torque command valueT_(e_max) based on the stator flux command value λ_(s_opt) in accordancewith Expression (36) to be described later. Expression (36) is apredetermined second calculation formula for upper limit torque commandvalue calculation. Expression (36) is a monomial obtained by multiplyingthe second-order term (λ_(s_opt) ²) of the stator flux command value bya predetermined coefficient. The predetermined coefficient is

$\frac{3}{4}\frac{P}{2}{\frac{L_{m}^{2}}{\sigma\; L_{s}^{2}L_{r}}.}$

The third block 13 c calculates a boundary speed ω_(e_c) in accordancewith Expression (34) to be described later. The boundary speed ω_(e_c)is a speed predetermined to be higher than a field weakening startingpoint, and determined by Expression (34). The field weakening startingpoint may be, for example, the rated speed of a motor.

The fourth block 13 d determines a speed region based on the powersupply angular frequency ω_(e) and the boundary speed ω_(e_c). The speedregion is divided into a normal operation region, Region I, and RegionII. Region I is a region between the boundary speed ω_(e_c) and thefield weakening starting point. Region II is a region higher than theboundary speed ω_(e_c).

Specifically, the fourth block 13 d determines the speed region inaccordance with the power supply angular frequency ω_(e) as follows. Thefield weakening starting speed is defined as ω_(base). Whenω_(e)<ω_(base), it is determined the speed is in the normal operationregion. When ω_(base)≤ω_(e)<ω_(e_c), it is determined that the speed isin Region I (a first field weakening region). When ω_(e_c)≤ω_(e), it isdetermined that the speed is in Region II (a second field weakeningregion). The fourth block 13 d outputs “0” when the current speed is inthe normal operation region or Region I. The fourth block 13 d outputs“1” when the current speed is in Region II.

When the output from the fourth block 13 d is “0”, the fifth block 13 eselects the upper limit torque command value T_(e_max) output from thefirst block 13 a. When the output from the fourth block 13 d is “I”, thefifth block 13 e selects the upper limit torque command value T_(e_max)output from the second block 13 b. The fifth block 13 e transmits theselected upper limit torque command value T_(e_max) to the sixth block13 f.

The sixth block 13 f determines a torque limiter range by using theupper limit torque command value T_(e_max) transmitted from the fifthblock 13 e. The torque limiter range is a range defined by an upperlimit torque value (+T_(e_max)) and a lower limit torque value(−T_(e_max)). The upper limit torque value is the upper limit torquecommand value T_(e_max). The lower limit torque value is a valueobtained by multiplying the upper limit torque value by a negativecoefficient. The negative coefficient is a predetermined value, and maybe “−1” or a value other than minus one. Also, instead of this negativecoefficient, the lower limit torque value may be set to zero bymultiplying zero by the upper limit torque value.

When the first torque command value T_(em1)* is in the torque limiterrange, the sixth block 13 f substitutes the first torque command valueT_(em1)* directly into the second torque command value T_(em)*. When thefirst torque command value T_(em1)* is larger than the upper limittorque value +T_(e_max), the sixth block 13 f substitutes the upperlimit torque value +T_(e_max) into the second torque command valueT_(em)*. When the first torque command value T_(em1)* is smaller thanthe lower limit torque value −T_(e_max), the sixth block 13 fsubstitutes the lower limit torque value −T_(e_max) into the secondtorque command value T_(em)*.

In this manner, the sixth block 13 f functions as a filter that allowsonly a torque command value in the torque limiter range to pass through.Accordingly, the sixth block 13 f generates the second torque commandvalue T_(em)*. The second torque command value T_(em)* is a valueobtained by correcting the first torque command value T_(em1)* so thatthe first torque command value T_(em1)* is limited to the torque limiterrange.

The appropriate flux command generation module 18 includes a first block18 a and a second block 18 b. The appropriate flux command generationmodule 18 generates the stator flux command value λ_(s_opt) inaccordance with the power supply angular frequency ω_(e).

The first block 18 a calculates the stator flux command value λ_(s_opt)in accordance with Expression (29) to be described later. The firstblock 18 a calculates a value obtained by dividing a stator voltagemaximum value V_(max) by the power supply angular frequency ω_(e).

The second block 18 b limits the stator flux command value λ_(s_opt)calculated by the first block 18 a to a certain range. The second block18 b employs a predetermined upper limiter flux value and limits thestator flux command value λ_(s_opt) to be equal to or smaller than theupper limiter flux value. The upper limiter flux value may be the ratedstator flux λ_(rate) as one of parameters of the motor in theembodiment.

The power supply angular frequency calculation module 19 calculates therotor electrical angular speed estimated value {circumflex over (ω)}_(r)by multiplying the rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r) calculated by the speed and phaseestimation module 21 by P/2. The value P is the pole number of the motor2. The power supply angular frequency calculation module 19 calculatesthe power supply angular frequency ω_(e) by adding the rotor mechanicalangular speed estimated value {circumflex over (ω)}_(rm-r) and a slipangular frequency estimated value {circumflex over (ω)}_(sl) outputtedfrom a first slip angular estimation module 32.

The slip angular frequency estimated value {circumflex over (ω)}_(sl)may be estimated by Expressions (43) or (44), as described later. In thefirst embodiment, the rotor dqs-axes flux estimated value λ_(qds) ^(S)outputted from the current/flux estimation module 20 and the rotordqs-axes current measured value i_(qds) ^(s) outputted from the secondthe second coordinate conversion module 17 are inputted to the firstslip angular estimation module 32. The first slip angular estimationmodule 32 performs a calculation based on Expression (43) to estimatethe slip angular frequency estimated value {circumflex over (ω)}_(sl).

Device Operation According to the First Embodiment

The following describes a theory for increasing an output torque in afield weakening region in DB-DTFC control. Expressions (1) to (4) areequations for an induction machine in an optional coordinate system.V _(qds) =R _(s) i _(qds) +jωλ _(qds) +pλ _(qds)  (1)0=R _(r) i _(qdr) +j(ω−ω_(r))λ_(qdr) +pλ _(qdr)  (2)λ_(qds) =L _(s) i _(qds) +L _(m) i _(qdr)  (3)λ_(qdr) =L _(r) i _(qdr) +L _(m) i _(qds)  (4)

A stator-side voltage equation in a synchronous coordinate system(synchronous with the power supply frequency) of the induction machinecan be given by Expression (5).V _(qds) ^(e) =R _(s) i _(qds) ^(e) +jω ^(e)λ_(qds) ^(e) +pλ _(qds)^(e)  (5)

Any differential term can be omitted in a steady state, and thusExpression (5) becomes Expression (6).V _(qds) ^(e) =R _(s) i _(qds) ^(e) +jω _(e)λ_(qds) ^(e)  (6)

The term of the power supply angular frequency ω_(e) is dominant in ahigh speed range, and thus finally, Expression (7) is obtained.V _(qds) ^(e) =jω _(e)λ_(qds) ^(e)  (7)

It is clear from Expression (7) that a voltage upper limit is reached asthe speed increases. In such a case, ω_(e) can be increased bydecreasing the flux. This is called field weakening.

One of features of the embodiment is a method of determining a fluxcommand for achieving an increased torque in the high frequency region.

1. Output Torque Increasing Technique

Studies performed by the inventor of the present application have founda novel torque output increasing technology, the contents of which willbe described below.

1.1 Stator Flux Limit with the Voltage and Current Limits Taken intoAccount

1.1.1 Restriction of Stator Voltage (Voltage Limit) by Maximum OutputVoltage of Inverter

The stator voltage maximum value V_(max) is limited by a DC voltageV_(dc) of the inverter and a modulation method. The stator voltagemaximum value V_(max) is V_(smax)=V_(dc)/2 for sinusoidal wave pulsewidth modulation (SPWM), or V_(smax)=V_(dc)/√{square root over (3)} forspace vector pulse width modulation (SVPWM). If necessary, in derivingthe maximum value Vsmax of the stator voltage, an influence of a forwardvoltage drop or a dead time of switching elements constituting theinverter 8 may be taken into consideration.

Another point is, if necessary, a coefficient slightly smaller than 1,for example 0.98, can be used to calculate Vsmax to account for thevoltage drop on the stator resistance.

Expression (7) gives a dqs-axes complex vector. Expression (8) is avoltage limit expression obtained from Expression (7).

$\begin{matrix}{{\left( \lambda_{qs}^{e} \right)^{2} + \left( \lambda_{d\; s}^{e} \right)^{2}} \leq \frac{V_{smax}^{2}}{\omega_{e}^{2}}} & (8)\end{matrix}$

FIG. 4 is a graph illustrating a stator flux locus for each power supply(operation) frequency with a voltage limit taken into account. A voltagelimit is illustrated as a circle in FIG. 4. The radius of each circle inFIG. 4 represents an allowable stator flux amplitude. The radius (upperlimit of the allowable stator flux amplitude) decreases as the powersupply (operation) angular frequency ω_(e) increases. The vertical axisexpresses the stator d-axis flux on general synchronous reference frameλ_(ds) ^(e), and the horizontal axis expresses the stator q-axis flux ongeneral synchronous reference frame λ_(qs) ^(e) in FIG. 10 from FIG. 4respectively.

1.1.2 Limit of Stator Current by Allowable Output Current of Inverter(Current Limit)

In addition to the voltage limit, another limit is placed on the statorflux amplitude. This limit is an allowable output current limit of theinverter 8 and allowable current through the motor

2. The Maximum Allowable Current I_(smax) is Expressed by Expression(9).(i _(qs) ^(e))²+(i _(ds) ^(e))² ≤I _(smax) ²  (9)

To use the stator flux in control, a relational expression between thestator current and the stator flux is derived. In the stationary state,Expression (2) as a voltage equation for the rotor becomes Expression(10).0=R _(r) i _(qdr) ^(e) +j(ω_(e)−ω_(r))κ_(qdr) ^(e)  (10)

When Expression (10) is rewritten for the q-axis and d-axis componentsseparately, Expressions (11) and (12) below are obtained.0=R _(r) i _(qr) ^(e)+(ω_(e)−ω_(r))λ_(dr) ^(e)  (11)0=R _(r) i _(dr) ^(e)+(ω_(e)−ω_(r))λ_(qr) ^(e)  (12)

Expressions (11) and (12) are simplified by performing coordinatetransformation on the rotor flux so that Expressions (13) and (14) areobtained.λ_(dr) ^(e)=λ_(r)  (13)λ_(qr) ^(e)=0  (14)

From Expressions (12) and (14), a rotor-side d-axis current on thesynchronous coordinate system is given by Expression (15).0=i _(dr) ^(e)  (15)

Expression (4) is rewritten in a manner divided for the q and d axes.λ_(qs) ^(e) =L _(s) i _(qs) ^(e) +L _(m) i _(qr) ^(e)  (16)λ_(ds) ^(e) =L _(s) i _(ds) ^(e) +L _(m) i _(dr) ^(e)  (17)

From Expressions (15) and (17), Relation Expression (18) between thestator current and the stator flux on the d-axis side can be obtained.λ_(ds) ^(e) =L _(s) i _(ds) ^(e)  (18)

Then, a relational expression between the stator current and the statorflux on the q-axis side is obtained. Expression (3) is rewritten in amanner divided for the d and q-axes.λ_(qr) ^(e) =L _(m) i _(qs) ^(e) +L _(r) i _(qr) ^(e)  (19)λ_(dr) ^(e) =L _(m) i _(ds) ^(e) +L _(r) i _(dr) ^(e)  (20)

Expression (21) can be obtained from Expressions (14) and (19).

$\begin{matrix}{i_{qr}^{e} = {{- \frac{L_{m}}{L_{r}}}i_{qs}^{e}}} & (21)\end{matrix}$

Lastly, the relational expression between the stator current and thestator flux on the q-axis side can be obtained from Expressions (16) and(21). In the expression, a represents a leakage coefficient listed in asymbol list.

$\begin{matrix}{\lambda_{qs}^{e} = {{\left( {L_{s} - \frac{L_{m}^{2}}{L_{r}}} \right)i_{qs}^{e}} = {\sigma\; L_{s}i_{qs}^{e}}}} & (22)\end{matrix}$

The current-voltage relation expressions (18) and (22) on the d-axis andq-axis sides are substituted into the current limit expression (9) tofinally obtain a current limit expression (23).

$\begin{matrix}{{\left( \frac{\lambda_{qs}^{e}}{\sigma\; L_{s}} \right)^{2} + \left( \frac{\lambda_{ds}^{e}}{L_{s}} \right)^{2}} \leq I_{smax}^{2}} & (23)\end{matrix}$

When illustrated on a stator flux plane, the expression depicts anellipse as illustrated in FIG. 5. FIG. 5 is a graph illustrating astator flux locus with a current limit taken into account.

1.1.3 Voltage and Current Limits

FIG. 6 is a graph illustrating stator flux loci with the voltage andcurrent limits taken into account. As described above in Sections 1.1and 1.2, the voltage and current limits of the inverter depict a circleand an ellipse, respectively, on the stator flux plane. These resultsindicate that, in theory, the stator flux exists in a hatched regioncommon to the voltage limit circle and the current limit ellipseillustrated in FIG. 6. Thus, to increase the output torque in the fieldweakening region, an appropriate combination of d-axis and q-axes fluxcommand vectors need to be selected from the hatched region.

1.2 Stator Flux for Achieving Increased Torque

As described in the previous section, a combination of (d-axis andq-axes) stator flux command vectors are limited by the current andvoltage limits of the inverter. Achievable torque differs between statorflux command vectors. The present section derives stator flux forachieving increased torque.

1.2.1 Torque Formula (Expression Only with Stator Flux)

A torque formula is expressed as Expression (24). This formula isexpressed only with the stator flux to depict torque on the dqs-axesplane of the stator flux.

$\begin{matrix}{T_{e} = {\frac{3}{2}\frac{P}{2}\frac{L_{m}}{\sigma\; L_{s}L_{r}}\left( {{\lambda_{qs}\lambda_{dr}} - {\lambda_{ds}\lambda_{qr}}} \right)}} & (24)\end{matrix}$

Expression (25) can be obtained from Expressions (15), (18), and (20).

$\begin{matrix}{\lambda_{dr}^{e} = {\frac{L_{m}}{L_{s}}\lambda_{ds}^{e}}} & (25)\end{matrix}$

Torque Formula (26) expressed only with the stator flux can be obtainedfrom Expressions (14), (24), and (25).

$\begin{matrix}{T_{e} = {\frac{3}{2}\frac{P}{2}\frac{L_{m}^{2}}{\sigma\; L_{s}^{2}L_{r}}\lambda_{qs}^{e}\lambda_{ds}^{e}}} & (26)\end{matrix}$

Expression (26) indicates that the stator flux locus depicts a hyperbolaon the dqs-axes flux plane when the torque is constant.

FIG. 7 is a graph illustrating voltage and current limits and torquecurves at a power supply frequency of 1.2 pu. FIG. 7 exemplarilyillustrates three torque curves having different sizes in addition tovoltage and current limits at a power supply frequency of 1.2 pu.

When the power supply frequency is 1.2 pu, the number of combinations ofstator fluxes satisfying the voltage and current limits is infinite forTL=1.0 pu. However, the number of such combinations is one for TL=1.7pu. This is a maximum torque achievable in theory. Both limits cannot besatisfied for TL=2.3 pu, and thus this torque cannot be achieved.

The radius of the voltage limit circle decreases as the power supplyfrequency increases. Accordingly, no torque curve exists in a regioncommon to the current limit ellipse and the voltage limit circle. Amaximum torque is achieved at a contact point between a torque curve andthe voltage limit circle. This point is a point of transition fromRegion I to Region II, which will be described later.

Specifically, in a particular region, the stator flux is limited by twolimits, namely, the current and voltage limits. In the other region,however, the stator flux is limited by one limit, namely, the voltagelimit. A flux command needs to be determined in each region due to thislimit difference.

The above-described two field weakening regions are referred to asRegion I and Region II. The following describes Region I, Region II, anda boundary point between these regions. The relation between each regionand the speed is listed below with description of terms.

ω_(e)<ω_(base): Normal operation region

ω_(base)≤ω_(e)<ω_(e_c): Region I (first field weakening region)

ω_(e_c)≤ω_(e): Region II (second field weakening region)

ω_(e_c): Boundary speed between Regions I and II

ω_(base): Field weakening start speed

$\omega_{base} = \frac{V_{smax}}{\lambda_{qds\_ rate}}$

The field weakening starting point ω_(base) is determined by the statorvoltage and the rated flux. In the first embodiment, it is assumed thatfield weakening automatically starts at voltage saturation.

1.2.2 Stator flux in Region I (ω_(base)<ω_(e)<ω_(e_c))

FIG. 8 is a graph illustrating a maximum torque curve with the voltageand current limits taken into account at different speeds in Region I.Region I corresponds to a case in which the power supply (operation)angular frequency ω_(e) exceeds a rated speed.

As illustrated in FIG. 8, when the operation frequency is not too high,an intersection point between the current limit ellipse and the voltagelimit circle exists above the straight line of λ_(ds) ^(e)=λ_(qs) ^(e).This operation frequency band is defined to be Region I.

In Region I, a stator flux for achieving the maximum torque is at theintersection point between the current limit ellipse and the voltagelimit circle. In FIG. 8, the maximum torque is achieved at operationpoint A for ω_(e)=1.2 pu and at operation point B for ω_(e)=1.8 pu.

Stator flux command vectors satisfying the intersection point are givenby Expressions (27) and (28) due to the voltage and current limits.

$\begin{matrix}{\lambda_{qs\_ opt}^{e} = \sqrt{\frac{{\sigma^{2}L_{s}^{2}I_{smax}^{2}} - \frac{\sigma^{2}V_{smax}^{2}}{\omega_{e}^{2}}}{1 - \sigma^{2}}}} & (27) \\{\lambda_{ds\_ opt}^{e} = \sqrt{\frac{\frac{V_{smax}^{2}}{\omega_{e^{2}}} - {\sigma^{2}L_{s}^{2}I_{smax}^{2}}}{1 - \sigma^{2}}}} & (28)\end{matrix}$

The amplitude of the stator flux is limited by Expression (29) as theradius of the voltage limit circle.

$\begin{matrix}{\lambda_{s\_ opt} = \frac{V_{smax}}{\omega_{e}}} & (29)\end{matrix}$

Expression (30), which calculates the maximum torque in the fieldweakening region I, is obtained from Expressions (26) to (29).

$\begin{matrix}{T_{e\_ max} = {{\frac{3}{2}\frac{P}{2}\frac{L_{m}^{2}}{\sigma\; L_{s}^{2}L_{r}}\lambda_{q{s\_ opt}}^{e}\lambda_{d{s\_ opt}}^{e}} = {\frac{3}{2}\frac{P}{2}\frac{L_{m}^{2}}{\left( {1 - \sigma^{2}} \right)L_{s}^{2}L_{r}}\sqrt{\left( {{L_{s}^{2}I_{smax}^{2}} - \lambda_{s\_ opt}^{2}} \right)\left( {\lambda_{s\_ opt}^{2} - {\sigma^{2}L_{s}^{2}I_{smax}^{2}}} \right)}}}} & (30)\end{matrix}$1.2.3 Stator flux (ω_(e_c)<ω_(e)) in Region II

The intersection point between the ellipse and the circle is below thestraight line of λ_(ds) ^(e)=λ_(qs) ^(e) at a higher speed. Nointersection point exists at a further higher speed. An operationfrequency band in which these phenomena occur is defined to be RegionII.

FIG. 9 illustrates a condition on the boundary speed between Regions Iand II. FIG. 9 is a graph indicating the condition on the boundary speedbetween Regions I and II. The boundary speed ω_(e_c) is calculated byExpressions (31) to (33) below.

Here, among the stator flux λ_(qds_c) ^(e), the λ_(qs_c) ^(e) representsthe q-axis component of the stator flux at the boundary speed, and theλ_(ds_c) ^(e) represents the d-axis component of the stator flux at theboundary speed.

$\begin{matrix}{{\left( \lambda_{q{s\_ c}}^{e} \right)^{2} + \left( \lambda_{d{s\_ c}}^{e} \right)^{2}} = \frac{V_{smax}^{2}}{\omega_{e\_ c}^{2}}} & (31) \\{{\left( \frac{\lambda_{q{s\_ c}}^{e}}{\sigma\; L_{s}} \right)^{2} + \left( \frac{\lambda_{d{s\_ c}}^{e}}{\; L_{s}} \right)^{2}} = I_{smax}^{2}} & (32) \\{\lambda_{q{s\_ c}}^{e} = \lambda_{d{s\_ c}}^{e}} & (33)\end{matrix}$

The boundary speed ω_(e_c) is finally obtained by Expression (34).

$\begin{matrix}{\omega_{e\_ c} = {\frac{V_{smax}}{\sigma\; L_{s}I_{smax}}\sqrt{\frac{1 + \sigma^{2}}{2}}}} & (34)\end{matrix}$

FIG. 10 is a graph illustrating maximum torque curves with the voltageand current limits taken into account at different power supply angularfrequencies in Region II. The maximum torque is achieved at points C andD illustrated in FIG. 10 for different operation speeds, respectively. Astator flux command vector for achieving the maximum torque in Region IIcan be obtained by Expression (35).

$\begin{matrix}{\lambda_{q{s\_ opt}}^{2} = {\lambda_{d{s\_ opt}}^{2} = \frac{V_{smax}}{\sqrt{2}\omega_{e}}}} & (35)\end{matrix}$

Similarly to Region I, the stator flux command amplitude is limited byExpression (29). The maximum torque achievable in the field weakeningregion H is expressed as Expression (36). Expression (36) is derivedfrom Expressions (26), (29), and (35).

$\begin{matrix}{T_{e\_ max} = {\frac{3}{4}\frac{P}{2}\frac{L_{m}^{2}}{\sigma\; L_{s}^{2}L_{r}}\lambda_{s\_ opt}^{2}}} & (36)\end{matrix}$

One of differences between Regions I and II is that the current isconstantly lower than the current limit (upper limit) due to a conditionon the maximum torque achievable in Region II. As a result, a statorflux amplitude for achieving the maximum torque is determined by theradius of the voltage limit circle in both of Regions I and IL

1.2.4 Control Block for Torque Increasing

The entire image of the above-described control is illustrated in FIGS.1 to 3. The stator flux command is generated by the first block 18 ainside the appropriate flux command generation module 18 in accordancewith Expression (29) based on the power supply frequency and the statorvoltage.

However, a certain limit is placed on the upper limit of the stator fluxcommand by the second block 18 b. An achievable torque command iscalculated by the torque command limit module 13 by using the statorflux command in accordance with Expressions (30) and (36). Theachievable torque differs between the field weakening region I and thefield weakening region II as described above.

According to the first embodiment above described, operation describedbelow is achieved.

The first block 18 a of the appropriate flux command generation module18 calculates the stator flux command value λ_(s_opt) to be smaller asthe power supply angular frequency ω_(e) increases.

The second block 18 b limits the stator flux command value λ_(s_opt) toa constant value at the rated flux (λ_(rate)) in a low speed range inwhich the power supply angular frequency ω_(e) is small. However, thesecond block 18 b places no limit in a high speed range in which thepower supply angular frequency ω_(e) is large. The second block 18 bplaces no limit at least in a speed region equal to or higher than thefield weakening starting point. Without the limit by the second block 18b, the stator flux command value λ_(s_opt) is calculated to be smalleras the power supply angular frequency ω_(e) increases.

As understood from Expressions (30) and (36), the upper limit torquecommand value T_(e_max) is calculated to be smaller as the stator fluxcommand value λ_(s_opt) is smaller. The width between the upper limittorque value (+T_(e_max)) and the lower limit torque value (−T_(e_max))decreases as the upper limit torque command value T_(e_max) is smaller.

The torque command limit module 13 can calculate the upper limit torquecommand value T_(e_max) to be smaller as the power supply angularfrequency ω_(e) increases in a somewhat high speed range. When the upperlimit torque command value T_(e_max) is calculated to be smaller, thewidth between the upper limit torque value (+T_(e_max)) and the lowerlimit torque value (−T_(e_max)) is set to be smaller.

Through such operation, the torque limiter range can be set to besmaller as the power supply angular frequency ω_(e) increases at leastin the field weakening regions I and II. An allowable torque range inwhich the control stability can be maintained tends to be smaller as thepower supply angular frequency ω_(e) increases. According to the firstembodiment, the torque limiter range is dynamically adjusted inaccordance with such a tendency.

Torque increase can be performed without degrading the control stabilitybecause a torque command value changes only in the torque limiter rangeadjusted to an appropriate range. Accordingly, when the motor isoperated fast in a field weakening region, the control stability and theoutput torque increase can be simultaneously achieved.

In the first embodiment, Expression (30) of the first block 13 a isapplied at a speed lower than the boundary speed ω_(e_c). The upperlimit torque command value T_(e_max) can be changed at an appropriatetendency in accordance with the second-order polynomial of the statorflux command value λ_(s_opt).

In the first embodiment, Expression (36) of the second block 13 b isapplied at a speed equal to or higher than the boundary speed ω_(e_c).The upper limit torque command value T_(e_max) can be appropriatelychanged in accordance with the monomial of the second-order term(λ_(s_opt) ²) of the stator flux command value at a tendency differentfrom that of Expression (30).

Accordingly, the upper limit torque command value T_(e_max) can bechanged at an appropriate tendency in each of the field weakeningregions I and II.

In the embodiment, the torque command limit module 13 includes the fifthblock 13 e. With this configuration, the upper limit torque commandvalue T_(e_max) calculated by the first block 13 a and the upper limittorque command value T_(e_max) calculated by the second block 13 b canbe selectively switched in accordance with the power supply angularfrequency ω_(e).

In the first embodiment, the fourth block 13 d is provided. With thisconfiguration, the first block 13 a, in other words, Expression (30) isapplied in both of Region I and the normal operation region.Accordingly, calculation of the upper limit torque command valueT_(e_max) can be seamlessly performed through the normal operationregion and the field weakening region.

In the first embodiment, the appropriate flux command generation module18 includes the second block 18 b. When Expression (29) is applied in alow speed range in which the power supply angular frequency ω_(e) issmall, the stator flux command value λ_(s_opt) is calculated to beexcessively large. The second block 18 b places an upper limiter toprevent the stator flux command value λ_(s_opt) from being calculated tobe excessively large.

Next, an example of the controller is described with the use of FIG. 20.FIG. 20 is a hardware configuration diagram of the controller for apower converter according to the embodiment 1 of the present invention.The hardware configuration in FIG. 20 may also be adapted to the secondembodiment and the third embodiment described later.

As illustrated in FIG. 20, each function of the controller 9 is executedby the processing circuit. The processing circuit includes a processor30 a and a memory 30 b.

For example, the processor 30 a is a central processing unit (CPU),e.g., a central processing device, a processing device, amicroprocessor, a microcomputer, a processor or a digital signalprocessor (DSP).

For example, the memory 30 b is a non-volatile or volatile semiconductormemory such as RAM, ROM, flash memory, EPROM, EEPROM, or magnetic disk,flexible disk, optical disk, compact disk, mini-disk or DVD.

In the processing circuit, a program stored in the memory 30 b isexecuted by the processor 30 a.

Second Embodiment Configuration of Device According to Second Embodiment

FIG. 11 is a configuration diagram of a motor system to which acontroller for a power convertor according to a second embodiment isapplied. The second embodiment differs from the first embodiment in thata motor system in the second embodiment is a sensor-less systemincluding no speed sensor, whereas the motor system according to thefirst embodiment is a sensor-equipped system including the speed sensor119.

In the second embodiment, the speed sensor 119 is omitted, and the speedand phase estimation module 21 is replaced with a speed and phaseestimation module 121. In addition, the first slip angular frequencyestimation module 32 is replaced by the second slip angular frequencyestimation module 33. Any other configuration is the same as that of thefirst embodiment.

The speed and phase estimation module 121 as a motor speed estimationmodule calculates the rotor electrical angle estimated value {circumflexover (θ)}_(r) and the rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r) based on the second torque command valueT_(em)*, a slip frequency estimated value ω_(sl) outputted from thesecond slip angular frequency estimation module 33, and the rotordqs-axes flux estimated value {circumflex over (λ)}_(qdr) ^(S).

The second slip angular frequency estimation module 33 uses the secondtorque command value T_(em)* and the rotor dqs-axes flux estimated value{circumflex over (λ)}_(qdr) ^(S) as input values to calculate the slipangular frequency estimation values {circumflex over (ω)}_(sl) based onEquation (44). The slip angular frequency estimated value {circumflexover (ω)}_(sl) is inputted to the speed/phase estimation module 121 andthe power supply angular frequency calculation module 19.

FIG. 12 is a block diagram of the main sections of the controller forthe power converter according to the embodiment 1 of the presentinvention. Next, the current/flux estimation module 20 and thespeed/phase estimation module 21 will be described with the use of FIG.12.

As illustrated in FIG. 12, the current/flux estimation module 20includes a current observation module 22, a first flux estimation module23 and a second flux estimation module 24.

The current observation module 22 calculates a stator dqs-axes currentestimated value î_(qds) ^(s) in a subsequent control period based on thevoltage command value V_(qds) ^(S)*, the stator dqs-axes currentmeasured value i_(qds) ^(s), a rotor mechanical angular speed estimatedvalue {circumflex over (ω)}_(rm-r) and a rotor dqs-axes flux estimatedvalue λ_(qdr) ^(S).

The stator dqs-axes current measured value i_(qds) ^(s) is the output ofthe second coordinate conversion module 17. The rotor mechanical angularspeed estimated value {circumflex over (ω)}_(rm-r) is an output of thespeed/phase estimation module 121. The rotor dqs-axes flux estimatedvalue {circumflex over (λ)}_(qdr) ^(S) is an output of the second fluxestimation module 24. The voltage command value V_(qds) ^(S)* is anoutput from the DB-DTFC calculation module 14.

On this occasion, the proportional gain K₃, the integral gain K₄, theestimated value {circumflex over (R)}_(eq) of the equivalent resistance,the estimated value {circumflex over (L)}_(r) of the rotor windinginductance, the estimated value {circumflex over (L)}_(m) of themagnetizing inductance, the rotor resistance {circumflex over (R)}_(r)and the imaginary number j, the rotor mechanical angular speed estimatedvalue {circumflex over (ω)}_(rm-r), the equivalent time constant τ_(eq),the control period T and the delay operator z⁻¹ are used.

Specifically, the current observation module 22 includes a first block22 a, a second block 22 b, a third block 22 c, a fourth block 22 d, afifth block 22 e, a sixth block 22 f, and a seventh block 22 g.

The first block 22 a calculates a value obtained by subtracting thestator dqs-axes current estimated value î_(qds) ^(s) from the statordqs-axes current measured value i_(qds) ^(s). The stator dqs-axescurrent estimated value î_(qds) ^(s) is the output of the seventh block22 g.

The output of the first block 22 a is inputted into the second block 22b and the third block 22 c. The second block 22 b is a proportionalcircuit of gain K₃. The third block 22 c is an integrating circuit ofgain K₄. The fourth block 22 d calculates a complemented value from arotor mechanical angular speed estimated value {circumflex over(ω)}_(rm-r) and a rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S). The rotor mechanical angular speed estimated value{circumflex over (ω)}_(rm-r) is an output of the speed/phase estimationmodule 121. The rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S) is an output of the second flux estimation module 24.

The fourth block 22 d calculates a value obtained by multiplying therotor dqs-axes flux estimated value {circumflex over (λ)}_(qdr) ^(S) bythe transfer coefficient G1 expressed by the following Expression (37).

$\begin{matrix}{{G\; 1} = {\frac{{\hat{L}}_{m}}{{\hat{L}}_{r}}\left( {\frac{{\hat{R}}_{r}}{{\hat{L}}_{r}} - {j{\hat{\omega}}_{{rm} - r}}} \right)}} & (37)\end{matrix}$

The fifth block 22 e calculates a value obtained by adding the voltagecommand value V_(qds) ^(S)*, the value calculated by the second block 22b, the value calculated by the third block 22 c, the value calculated bythe fourth block 22 d.

The sixth block 22 f calculates a value obtained by dividing the valuecalculated by the fifth block 22 e by the estimated value {circumflexover (R)}_(eq) of the equivalent resistance.

The output of the sixth block 22 f is input into the seventh block 22 g.The seventh block 22 g calculates a transfer function expressed in theExpression (38). The seventh block 22 g outputs a stator dqs-axescurrent estimated value î_(qds).

$\begin{matrix}{{G\; 2} = \frac{z^{- 1}\left( {1 - e^{{- T}/\tau_{eq}}} \right)}{1 - {z^{- 1}e^{{- T}/\tau_{eq}}}}} & (38)\end{matrix}$

The stator dqs-axes current estimated value î_(qds) ^(s) is input intothe first block 22 a and the second flux estimation module 24.

The first flux estimation module 23 calculates a rotor dqs-axes fluxestimated value λ_(qdr) ^(S)* based on the rotor electrical angleestimated value {circumflex over (θ)}_(r) and the stator dqs-axescurrent measured value i_(qds) ^(s). The rotor electrical angleestimated value {circumflex over (θ)}_(r) is an output of thespeed/phase estimation module 121. The stator dqs-axes current measuredvalue i_(qds) ^(s) is an output of the second coordinate conversionmodule 17.

On this occasion, the estimated value {circumflex over (L)}_(m) of themagnetizing inductance of the motor 2, the rotor time constant τ_(r) ofthe motor 2, the control period T and the delay operator z⁻¹ are used.

Specifically, the first flux estimation module 23 includes a first block23 a, a second block 23 b, and a third block 23 c.

The first block 23 a converts the stator dqs-axes current measured valuei_(qds) ^(s) to the value of the rotor coordinate system by the rotorelectrical angle estimated value {circumflex over (θ)}_(r).

The output of the first block 23 a is the input of the second block 23b.

The second block 23 b multiplies the value calculated by the first block23 a by the transfer function G3 expressed by the following Expression(39) to calculate the stator dqs-axes flux estimated value of theprimary hold.

$\begin{matrix}{{G\; 3} = \frac{{\hat{L}}_{m}{{\left( {1 - {\tau_{r}\text{/}T} + {\tau_{r}\text{/}T\; e^{{- T}/\tau_{r}}}} \right) + {\left( {\frac{\tau_{r}}{T} - {\tau_{r}\text{/}T\; e^{{- T}/\tau^{r}}} - e^{{- T}/\tau_{r}}} \right)z^{- 1}}}}}{1 - {\left( e^{{- T}/\tau_{r}} \right)z^{- 1}}}} & (39)\end{matrix}$

The output of the second block 23 b is the input of the third block 23c.

The third block 23 c converts the output of the second block 23 b to thevalue of the rotor coordinate system by the rotor electrical angleestimated value {circumflex over (θ)}_(r). The output of the secondblock 23 b is the rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S)*. The rotor dqs-axes flux estimated value λ_(qdr) ^(S)*is inputted into second flux estimation module 24.

The second flux estimation module 24 calculates a stator dqs-axes fluxestimated value {circumflex over (λ)}_(qds) ^(S) and a rotor dqs-axesflux estimated value {circumflex over (λ)}_(qdr) ^(S) in a subsequentcontrol period based on the voltage command value V_(qds) ^(S)*outputted from the DB-DTFC calculation module 14, the stator dqs-axescurrent measured value i_(qds) ^(s) output of the second coordinateconversion module 17, the stator dqs-axes current estimated valueî_(qds) ^(s) estimated by the current observation module 22, the rotordqs-axes flux estimated value λ_(qdr) ^(S)* estimated by the first fluxestimation module 23.

On this occasion, the proportional gain K₁, the integral gain K₂, theestimated value {circumflex over (R)}_(s) of the stator resistance, theleakage factor σ, the estimated value {circumflex over (L)}_(r) of therotor winding inductance, the estimated value of {circumflex over(L)}_(s) of the stator winding inductance, the estimated value{circumflex over (L)}_(m) of the magnetizing inductance, the controlperiod T and the delay operator z⁻¹ are used.

Specifically, the second flux estimation module 24 includes a firstblock 24 a, a second block 24 b, a third block 24 c, a fourth block 24d, a fifth block 24 e, a sixth block 24 f, a seventh block 24 g, and aneighth block 24 h, a ninth block 24 i, and a tenth block 24 j.

The first block 24 a calculates a value of the voltage drop bymultiplying the stator dqs-axes current measured value i_(qds) ^(s) bythe estimated value {circumflex over (R)}_(s) of the stator resistance.

The second block 24 b subtracts the rotor dqs-axes flux estimated value{circumflex over (λ)}_(qdr) ^(S) which is the output of the tenth block24 j from the rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S)*.

The third block 24 c calculates a value obtained by subtracting thevalue calculated by the first block 24 a from the voltage command valueV_(qds) ^(S)*.

The fourth block 24 d, the fifth block 24 e, and the sixth block 24 ffunction as a transition frequency determination unit 24 k. Thetransition frequency determining unit 24 k determines the transitionfrequency between the first flux estimation module 23 and the secondflux estimation module 24.

Specifically, the fourth block 24 d calculates a value obtained bymultiplying the value calculated by the second block 24 b by theproportional gain K₁.

The fifth block 24 e is an integrating circuit of gain K₂. The fifthblock 24 e calculates an integral of the output of the fourth block 24d.

The sixth block 24 f calculates a value of the input voltage of themotor 2 by adding the value calculated by the third block 24 c, thevalue calculated by the fourth block 24 d and the value calculated bythe fifth block 24 e.

The seventh block 24 g calculates the stator dqs-axes flux estimatedvalue {circumflex over (λ)}_(qds) ^(S) by integrating the output of thesixth block 24 f.

The eighth block 24 h calculates a value obtained by multiplying thestator dqs-axes current estimated value î_(qds) ^(s) by the factor G4expressed by the following Expression (40).G4=σ{circumflex over (L)} _(s)  (40)

The ninth block 24 i calculates a value obtained by subtracting thevalue calculated by the eighth block 24 h from the stator dqs-axes fluxestimated value {circumflex over (λ)}_(qds) ^(S) calculated by theseventh block 24 g.

The tenth block 24 j multiplies the value calculated by the ninth block24 i by the factor G5 expressed by the following Expression (41) tocalculate the rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S).

$\begin{matrix}{{G\; 5} = \frac{{\hat{L}}_{r}}{{\hat{L}}_{m}}} & (41)\end{matrix}$

For example, in the case where the frequency of the rotor flux of themotor 2 is lower than the transition frequency, the first fluxestimation module 23 is dominant. For example, in the case where thefrequency of the rotor flux of the motor 2 is higher than the transitionfrequency, the second flux estimation module 24 is dominant. As aresult, the rotor dqs-axes flux estimated value is accuratelycalculated.

The speed/phase estimation module 21 includes a phase estimation module25, a slip angle estimation module 26, a flux vector rotation module 27,a phase error estimation module 28 and a speed and position observingmodule 29.

The phase estimation module 25 calculates an estimated value e^(jθ) ^(e)of a phase of the flux vector of the motor 2 from the rotor dqs-axesflux estimated value {circumflex over (λ)}_(qdr) ^(S) calculated by thecurrent/flux estimation module 20. For example, the phase of flux vectorcan be the phase of the stator flux vector calculated by thecurrent/flux estimation module 20. The rotor dqs-axes flux estimatedvalue {circumflex over (λ)}_(qdr) ^(S) which is the output of thecurrent/flux estimation module 20 is input into the phase estimationmodule 25.

In particular, the phase estimation module 25 calculates an estimatedvalue e^(jθ) ^(e) of a phase of the stator flux vector of the motor 2 bythe following Expression (42).

$\begin{matrix}{e^{j\;\theta_{e}} = {{{\cos\;\theta_{e}} + {j\;\sin\;\theta_{e}}} = {\frac{{\hat{\lambda}}_{q*}^{s}}{{\hat{\lambda}}_{{qd}*}^{s}} + \frac{{\hat{\lambda}}_{d*}^{s}}{{\hat{\lambda}}_{{qd}*}^{s}}}}} & (42)\end{matrix}$

Although the power phase can be calculated by using any one of thestator flux {circumflex over (λ)}_(qds) and the rotor flux {circumflexover (λ)}_(qdr) in the Expression (42), the stator flux is used for theestimation in the first embodiment. In the Expression (42), the suffix“*”, i.e. an asterisk, is “s” when a calculation is done using thestator flux {circumflex over (λ)}_(qds). On the other hand, in theExpression (42), the suffix “*” is “r” when a calculation is done usingthe rotor flux {circumflex over (λ)}_(qdr).

An estimated value {circumflex over (ω)}_(sl) of a slip frequency of themotor 2 is calculated by the second slip angular frequency estimationmodule 33. The estimated value {circumflex over (ω)}_(sl) of the slipfrequency is inputted to the slip angle estimation module 26.

The slip angle estimation module 26 calculates a slip angle estimatedvalue {circumflex over (θ)}_(sl). The slip angle estimated value{circumflex over (θ)}_(sl) is calculated by integrating the estimatedvalue {circumflex over (ω)}_(sl) of the slip frequency. The sine andcosine of the slip angle estimated value {circumflex over (θ)}_(sl) iscalculated by the sine/cosine calculation unit 27 a and is inputted tothe flux vector rotation module 27.

The flux vector rotation module 27 calculates a first rotor electricalangle estimated value {circumflex over (θ)}_(r1) based on the estimatedvalue e^(jθ) ^(e) of the phase of the flux vector calculated by thephase estimation module 25 and the output of the sine/cosine calculationmodule 27 a.

The sine and cosine of the rotor electric angle estimated value{circumflex over (θ)}_(r) is calculated by the sine/cosine calculationunit 28 a. The calculated values of the sine and cosine of the rotorelectric angle estimated value {circumflex over (θ)}_(r) are inputted tothe phase error estimation module 28.

The phase error estimation module 28 performs additive theorem operationby using the two rotor electrical angles calculated by the flux vectorrotation module 27 and the speed and position observing module 29. Theestimated value {circumflex over (θ)}_(err) of rotor electrical angleerror is calculated by using approximation of sin Δθ≈Δθ when Δθ isminute.

The speed and position observing module 29 calculates rotor mechanicalangular estimated values {circumflex over (ω)}_(rm-r) and {circumflexover (ω)}_(rm-r), a mechanical angle estimated value {circumflex over(θ)}_(rm), electrical angle estimated value {circumflex over (θ)}_(r)based on the estimated value {circumflex over (θ)}_(err) calculated byphase error estimation module 28.

Specifically, the speed and position observing module 29 includes afirst block 29 a, a second block 29 b, a third block 29 c, a fourthblock 29 d, a fifth block 29 e, a seventh block 29 f, an eighth block 29g, and a ninth block 29 h and a tenth block 29 i.

The first block 29 a is an integrating circuit of gain K_(io). The firstblock 29 a calculates an integral of the estimated value {circumflexover (θ)}_(err) calculated by phase error estimation module 28. Thesecond block 29 b corrects the estimated value {circumflex over(θ)}_(err) estimated by the phase error estimation module 28 bymultiplying the estimated value {circumflex over (θ)}_(err) estimated bythe phase error estimation module 28 and the gain K_(so).

The third block 29 c multiples the estimated value {circumflex over(θ)}_(err) estimated by the phase error estimation module 28 and thegain b_(o).

The fourth block 29 d calculates a value obtained by adding the valuecalculated by the first block 29 a and the value calculated by thesecond block 29 b.

The fifth block 29 e calculates a value obtained by adding the valuecalculated by the fourth block 28 d and the torque command valueT_(em)*, and outputs the calculated value to the sixth block 29 f. Thesixth block 29 f divides the input value by the estimated value Ĵ_(p) ofmoment of inertia and outputs it to the seventh block 29 g.

The seventh block 29 g calculates a rotor mechanical angular speedestimated value {circumflex over (ω)}_(rm-l). The seventh block 29 g isan integrating circuit. The seventh block 29 g calculates an integral ofthe output of the sixth block 29 f.

The eighth block 29 h calculates the speed correction value by dividingthe value calculated by the third block 29 c by the estimated valueĴ_(p) of the moment of inertia.

The ninth block 29 i calculates a rotor mechanical angular speedestimated value {circumflex over (ω)}_(rm-r) by adding the rotormechanical angular speed estimated value {circumflex over (ω)}_(rm-l)calculated by the seventh block 29 g and the speed correction valueobtained by the eighth block 29 h.

The tenth block 29 j calculates a rotor mechanical angle estimated value{circumflex over (θ)}_(rm). The tenth block 29 j is an integratingcircuit. The seventh block 29 g calculates an integral of the rotormechanical angular speed estimated value {circumflex over (ω)}_(rm-r)

The eleventh block 29 k calculates the rotor electrical angle estimatedvalue {circumflex over (θ)}_(r) by dividing the value obtained bymultiplying the rotor mechanical angle estimated value {circumflex over(θ)}_(rm) calculated by the tenth block 29 j by the number of poles P ofthe motor 2 by 2.

Operation of Device According to Second Embodiment

1. Field Weakening Region Increase in Self-Sensing

The system according to the second embodiment is a sensor-less systemincluding no speed sensor, and has a self-estimation function ofestimating, for example, a speed by using various parameters in thesystem. The self-estimation function is also referred to as“self-sensing”.

As described above in the first embodiment, the output torque can beincreased by selecting a combination of stator flux command vectors in afield weakening region. This allows fast operation when the load torqueis large, and accordingly, leads to increase of the field weakeningregion. The present section describes discussions related to the fieldweakening region increase by the self-sensing.

In the system according to the second embodiment illustrated in FIGS. 11and 12, a flux observer is constituted by the first flux estimationmodule 23 and the second flux estimation module 24. The speed and phaseestimation module 121 performs speed estimation by using a flux observerfunction of the current and flux estimation module 20. This speedestimation using the flux observer function can be performed by usingone of a stator flux estimated value and a rotor flux estimated value.

In addition, a slip angular frequency is input to the speed and phaseestimation module 121. Estimation of the slipping angular frequency canbe performed by using one of the stator flux estimated value and therotor flux estimated value. Thus, either the stator flux estimated valueor the rotor flux estimated value can be optionally selected as an inputvalue used in the speed estimation and the slipping angular frequencyestimation.

The speed and position observation module 29 in the speed and phaseestimation module 121 illustrated in FIG. 12 is equivalent to the motionobserver.

The slipping angular frequency calculated based on the stator flux isexpressed by Expression (43). The slipping angular frequency calculatedbased on the rotor flux is expressed by Expression (44).

When the power supply phase e^(jθ) ^(e) is estimated by using the statorflux, the slip calculation is performed by the Expression (43). On theother hand, when power supply phase e^(jθ) ^(e) is estimated by usingthe rotor flux, the slip calculation is performed by the Expression(44). One of {circumflex over (ω)}_(sl-s) and {circumflex over(ω)}_(sl-r) is eventually used, and the eventually used value isexpressed as {circumflex over (ω)}_(sl).

It should be noted that, in the Expression (43), a superscript symbol“es” of each sign represents a synchronous coordinate system developedwith the stator flux as a reference. The stator flux value λ_(qs) ^(s)of the stationary coordinate system is a sine wave signal, and thedevelopment is done based on the stator flux as a reference.

In the synchronous coordinate system, the origin of the coordinatesystem is reset in each control cycle so as to ensure λ_(qs) ^(es)=0.Thereby, other signals are also converted accordingly. The advantageouseffect of this coordinate transformation is that specific calculationscan be easier.

$\begin{matrix}{{\hat{\omega}}_{sl\_ s} = \frac{L_{s}\left\lbrack {i_{qs}^{es} + {\sigma\;\tau_{r}\frac{d\; i_{qs}^{es}}{d\; t}}} \right\rbrack}{\tau_{r}\left( {\lambda_{ds}^{es} - {\sigma\; L_{s}i_{ds}^{es}}} \right)}} & (43) \\{{\hat{\omega}}_{sl\_ r} = {\frac{4}{3\; P}\frac{T_{em}^{*}}{{\lambda_{qdr}^{s}}^{2}}R_{r}}} & (44)\end{matrix}$2. Low Pass Filter Effect

The rotor flux has less noise than the stator flux. This is because therotor flux is less affected by a high frequency component included involtage due to PWM. This will be described below by using Expression(45) as a stator flux equation and Expression (46) as a rotor fluxequation.

$\begin{matrix}{{p\;\lambda_{qds}^{s}} = {V_{qds}^{s} - {\frac{R_{s}}{\sigma\; L_{s}}\lambda_{qds}^{s}} + {\frac{R_{s}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qdr}^{s}}}} & (45) \\{{p\;\lambda_{qdr}^{s}} = {{\frac{R_{r}L_{m}}{\sigma\; L_{s}L_{r}}\lambda_{qds}^{s}} - {\left( {\frac{R_{r}}{\sigma\; L_{r}} - {j\;\omega_{r}}} \right)\lambda_{qdr}^{s}}}} & (46)\end{matrix}$

The stator flux receives an input of voltage as indicated in Expression(45). Thus, the stator flux is directly affected by a high frequencycomponent of voltage due to PWM. However, this influence on the rotorflux is reduced by a filter. This filter effect will be described below.

Expression (46) is provided with Laplace transform and rewritten byreplacing coefficients of the stator flux and the rotor flux on theright hand side with A and B, respectively.

p λ_(qdr)^(s) = A λ_(qds)^(s) − B λ_(qdr)^(s)$\lambda_{qdr}^{s} = {\frac{1}{s}\left( {{A\;\lambda_{qds}^{s}} - {B\;\lambda_{qdr}^{s}}} \right)}$${\left( {1 + \frac{B}{s}} \right)\lambda_{qdr}^{s}} = {\frac{A}{s}\lambda_{qds}^{s}}$$\lambda_{qdr}^{s} = {\frac{s}{s + B}\frac{A}{s}\lambda_{qds}^{s}}$

Finally, Expression (47) below is derived.

$\begin{matrix}{\lambda_{qdr}^{s} = {\frac{A}{s + B}\lambda_{qds}^{s}}} & (47)\end{matrix}$

As understood from Expression (47), the rotor flux is equivalent to fluxobtained by applying a low pass filter to the stator flux. The rotorflux is a value obtained based on the stator flux, and the stator fluxis affected by a high frequency component of voltage. However, influenceof voltage noise on the rotor flux is reduced by the above-described lowpass filter effect.

When the flux amplitude is reduced by field weakening, the S/N ratiodegrades. Additionally, in self-sensing in which the stator flux is usedin phase estimation, control is likely to be unstable as compared to acase in which the rotor flux is used. This is because the stator flux isdirectly affected by a high frequency component of voltage.

The above-described low pass filter effect is achieved by the eighthblock 24 h, the ninth block 24 i, and the tenth block 24 j, inparticular, among the blocks of the second flux estimation module 24illustrated in FIG. 12.

4. Test Results

FIGS. 13 to 15 are diagrams showing an example of the test results offield weakening in the second embodiment. The response when the speedcommand is increased from 0.1 pu to 2.0 pu is shown. In both figures,the horizontal axis is the time axis.

The vertical axis in FIG. 12 represents the angular speed command valueω_(rm)* of the motor 2 and the measured speed value ω _(rm). Thevertical axis in FIG. 14 represents the estimated torque of the motor 2and the torque limit value. The vertical axis in FIG. 15 represents theamplitude of the stator flux command value and the estimated stator fluxamplitude.

The estimated torque is calculated by multiplying the stator fluxestimated value and the rotor flux estimated value. As shown in FIG. 13,the motor 2 is stably accelerated to a speed of 2.0 PU. According toFIG. 14, the estimated torque of the motor 2 is equal to the torquelimit value during acceleration, but when the acceleration is completed,the estimated torque of the motor 2 is smaller than the torque limitvalue.

This performance depends on a bandwidth obtained by internalproportional integral gain of each observer. Suppressing the bandwidthat relatively low level makes it possible that a high frequency bandnoise component is not included in the estimated speed.

In the field weakening operation, it is preferable to set the bandwidthof the speed and phase estimation module 121 to be relatively low toreduce noise included in the estimated speed. A speed (phase) forself-sensing can be calculated from various waveforms of current,voltage, and flux, for example.

The second embodiment above described includes at least twocharacteristic configurations.

The first characteristic configuration is the speed and phase estimationmodule 121. This configuration achieves a sensor-less system equivalentto the system according to the first embodiment. The sensor-lessconfiguration provides a technological advantage such as the torquelimit adjustment in a field weakening region achieved according to thefirst embodiment, and allows any speed sensor to be omitted.

The second characteristic configuration is the usage of the rotordqs-axes flux estimated value {circumflex over (λ)}_(qdr) ^(S) as aninput value to the speed and phase estimation module 121. The speed andphase estimation module 121 calculates an estimated value of a speed ofthe motor in a subsequent control period based on the rotor dqs-axesflux estimated value {circumflex over (λ)}_(qdr) ^(S).

A rotor flux estimated value contains less noise than a stator fluxestimated value, which leads to highly accurate calculation of theestimated speed value. As a result, the effect of increasing a fieldweakening region can be achieved in the self-sensing of the sensor-lesssystem.

Modification of Second Embodiment

FIG. 16 is a configuration diagram of a motor system to which acontroller for a power convertor according to a modification of thesecond embodiment is applied. In the modification illustrated in FIG.16, a switching block 122 is provided between the speed and phaseestimation module 121 and the current and flux estimation module 20. Inthe modification shown in FIG. 16, the first slip angular frequencyestimation module 32, the second slip angular frequency estimationmodule 33, and a switching block 123 connected thereto are provided.

The stator dqs-axes flux estimated value {circumflex over (λ)}_(qds)^(S) outputted from the current and flux estimation module 20 and thestator dqs-axes current measured value i_(qds) ^(s) outputted from thesecond coordinate conversion module 17 are inputted to the first slipangular frequency estimation module 32. The first slip angular frequencyestimation module 32 performs a calculation based on Expression (43) byusing the above inputted values to calculate the slip angular frequencyestimated value {circumflex over (ω)}_(sl_s).

The second torque command value T_(em)* and the rotor dqs-axes fluxestimated value {circumflex over (λ)}_(qdr) ^(S) are inputted to thesecond slip angular frequency estimation module 33.

The second slip angular frequency estimation module 33 performs acalculation based on Expression (44) by using the above inputted valuesto calculate the slip angular frequency estimated value {circumflex over(ω)}_(sl_r).

The switching block 122 selectively transmits one of the estimated valueselected from the rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S) and the stator dqs-axes flux estimated value {circumflexover (λ)}_(qds) ^(S) to the speed and phase estimation module 121.

The switching block 123 selectively transmits one of the estimated valueselected from the slip angular frequency estimated value {circumflexover (ω)}_(sl_s) outputted from the first slip angular frequencyestimation module 32 and the slip angular frequency estimated value{circumflex over (ω)}_(sl_r) outputted from the second slip angularfrequency estimation module 33 to the speed and phase estimation module121.

The switching block 122 works together with the switching block 123.

When the switching block 122 selects the rotor dqs-axes flux estimatedvalue {circumflex over (λ)}_(qdr) ^(S), the switching block 123 selectsthe slip angular frequency estimated value {circumflex over (ω)}_(sl_r)outputted from the second slip angular frequency estimation module 33.

On the other hand, when the switching block 122 selects the statordqs-axes flux estimated value {circumflex over (λ)}_(qds) ^(S), theswitching block 123 selects the slip angular frequency estimated value{circumflex over (ω)}_(sl_s) outputted from the first slip angularfrequency estimation module 32.

The criterion by which the switching block 122 and the switching block123 switch the above estimated values may be predetermined.

For example, the rotor dqs-axes flux estimated value {circumflex over(λ)}_(qdr) ^(S) may be selected if the power supply angular frequencyω_(e) is equal to or higher than a predetermined value. For example, thestator dqs-axes flux estimated value {circumflex over (λ)}_(qds) ^(S)may be selected if the power supply angular frequency ω_(e) is lowerthan the predetermined value.

FIG. 17 is a configuration diagram of the controller for a powerconvertor according to the modification of the second embodiment. Theconfiguration illustrated in FIG. 17 does not include theabove-described second characteristic configuration included in thesecond embodiment. Accordingly, in the modification illustrated in FIG.17, the stator dq-axis flux estimated value {circumflex over (λ)}_(qds)^(S) is an input value to the speed and phase estimation module 121.

The modification illustrated in FIG. 17 cannot provide theabove-described effects of “improvement of calculation accuracy of theestimated speed value and sensor-less field weakening region increase bythe use of the rotor flux”. However, even though those effects are notobtained, the modification illustrated in FIG. 17 has advantages of“torque increasing same as that of the first embodiment” and “no need toprovide a speed sensor in a sensor-less system”.

The controller 11 according to the second embodiment may be achieved byusing the structure illustrated in the hardware configuration diagram inFIG. 20.

FIG. 21 is a graph of relative RMS noise as flux decreases. Asillustrated in FIG. 21, relative RMS error included in the estimatedspeed increases as the flux decreases. This is because that the fluxamplitude decreases as the rotor speed increases while the amplitudes ofa PWM harmonic wave and a noise of current measurements are constant.

The noise is normalized to 1 when the flux is 1 pu. The noise increasesto 13 approximately when the flux becomes near 0.1 pu.

The formula of the noise is rms ({circumflex over (ω)}_(r)−{circumflexover (ω)}_(avg)). In the formula, {circumflex over (ω)}_(r) representsan instantaneous speed at each sampling point, and {circumflex over(ω)}_(avg) represents an average speed between sampling points.

It is preferable to reduce influence of noise at low amplitude flux(that is, in a high speed range). To achieve this, it is preferable toset controller gains bo, Kso, and Kio to be low when rotor flux basespeed estimation is performed by the speed and phase estimation modules21, 121 illustrated in FIGS. 3, 12 and 17.

In FIGS. 12 and 17, the controller gain bo is the gain of a third block29 c. The controller gain Kso is the gain of a second block 29 b. Thecontroller gain Kio is the gain of a first block 29 a. The controllergains bo, Kso, and Kio achieve a function as a filter.

Decrease of the controller gains bo, Kso, and Kio is equivalent todecrease of the cutoff frequency of the filter. Accordingly, noise in ahigh frequency component included in a rotor flux input of an observercan be reduced.

Third Embodiment

FIG. 18 is a configuration diagram of a motor system to which acontroller for a power convertor according to a third embodiment isapplied.

The torque command limit module 13, the appropriate flux commandgeneration module 18, and the power supply angular frequency calculationmodule 19, which are provided in the first and second embodiments, areomitted in the motor system according to the third embodiment. Any otherconfiguration is the same as that of the second embodiment.

A value input to the DB-DTFC calculation module 14 is changed due to theconfiguration omission. In place of the second torque command valueT_(em)*, the first torque command value T_(em1)* calculated by the speedcontrol module 12 is input to the DB-DTFC calculation module 14. Inplace of the stator flux command value λ_(s_opt) generated by theappropriate flux command generation module 18, the rated stator fluxλ_(rate) is input to the DB-DTFC calculation module 14.

Since the torque command limit module 13 and the like are omitted, theadvantage of “operation at high power in a wider frequency range”, whichis achieved in the first embodiment, is restricted in the thirdembodiment as compared with the first embodiment.

However, similarly to the second embodiment, the speed and phaseestimation module 121 is provided and an input value to the speed andphase estimation module 121 is the rotor dqs-axes flux estimated value{circumflex over (λ)}_(qdr) ^(S) in the third embodiment. Thus, theeffects of “improvement of calculation accuracy of the estimated speedvalue and sensor-less field weakening region increase by the use of therotor flux”, which are the same as those of the second embodiment, canbe obtained in the third embodiment.

Modification of Third Embodiment

FIG. 19 is a configuration diagram of a motor system to which acontroller for a power convertor according to a modification of thethird embodiment is applied.

The configuration shown in FIG. 18 is configured with the dead-beatdirect torque & flux control (DB-DTFC) method which does not use a PIcurrent controller to generate a voltage command.

However, the configuration in the third embodiment can be implementednot only in the DB-DTFC method but also in a Field Oriented Controlmethod. Field Oriented Control method is also referred to as FOC methodhereinafter.

The FOC method is a method in which a current component for generating atorque (rotational force) and a current component for generating a fluxare separated from each other, and each current component isindependently controlled as a direct current amount.

One of FOC methods is DFOC (Direct Field Oriented Control) method. DFOCmethod is a method of directly estimating and controlling the fluxvector by a flux sensor or a flux observer. FIG. 19 illustrates aconfiguration diagram of a motor system to which a controller, which isconfigured based on DFOC method, for a power convertor according to amodification of the third embodiment is applied.

Another one of the FOC methods is IFOC (Indirect Field Oriented Control)method. The IFOC method uses indirect type vector control (also calledslip frequency type vector control) that controls a slip of an inductionmachine regardless of flux estimation or flux detection.

In the modification shown in FIG. 19, DB-DTFC calculation module 14 isreplaced with a DFOC calculation module 314, and the first coordinateconversion module 15 is replaced with a fifth coordinate conversionmodule 15 a.

The controller 11 includes a third coordinate conversion module 17 a anda fourth coordinate conversion module 17 b. The v-phase stator currentI_(vs) and the w-phase stator current I_(ws) are inputted to the thirdcoordinate conversion module 17 a.

The three phase stator voltage command value V_(us)*, V_(vs)*, andV_(ws)*, which is outputted from the fifth coordinate conversion module15 a, is inputted to the fourth coordinate conversion module 17 b.

The first torque command value T_(em1)* calculated by the speed controlmodule 12 is inputted to the DFOC calculation module 314 in the systemaccording to DFOC method shown in FIG. 19.

With respect to a flux command, although the rated flux value λ_(rate)is inputted to the DFOC calculation module 314 as the stator fluxcommand value λ_(s)*, other system based on DFOC method may perform acontrol based on the rotor flux command value {circumflex over(λ)}_(qdr) ^(S)* instead of the stator flux command value λ_(s)*. Sincethe rotor flux command value {circumflex over (λ)}_(qdr) ^(S)* can becalculated from of the stator flux command value λ_(s)*, only the statorflux command value λ_(s)* is illustrated in FIG. 19.

The speed and phase estimation module 121 inputs the stator power supplyphase estimated value {circumflex over (θ)}_(e) as a reference signal tothe third coordinate conversion module 17 a and the fifth coordinateconversion module 15 a.

On the basis of the reference signal, the third coordinate conversionmodule 17 a converts the input signal to a signal of the rotatingcoordinate system of γ and δ components orthogonal to each other.

By appropriately selecting the phase of the reference signal, it ispossible to make the γ component in-phase with the reference and to makethe 8 component orthogonal to the reference.

The third coordinate conversion module 17 a performs a coordinateconversion to obtain a γ-axis stator current i_(γ) and a δ-axis statorcurrent i_(δ), and outputs the current values to the DFOC calculationmodule 314.

Conversely to the above, the fifth coordinate conversion module 15 aconverts the voltage command value V_(γ)* of the γ component and thevoltage command value V_(δ)* of the δ component, which are outputs fromthe DFOC calculation module 314, to the three-phase stator voltagecommand values V_(us)*, V_(vs)*, and V_(ws)*. The output from the fifthcoordinate conversion module 15 a is inputted to the PWM control module16 and the fourth coordinate conversion module 17 b.

The fourth coordinate conversion module 17 b converts the inputtedthree-phase stator voltage command values V_(us)*, V_(vs)*, and V_(ws)*which are values in fixed coordinate system to the voltage command valueV_(qds) ^(S)* of two-axes components of dqs-axes. The converted valuesby the fourth coordinate conversion module 17 b are inputted to thecurrent and flux estimation module 20.

The stator flux estimated value {circumflex over (λ)}_(qds) ^(s) and therotor flux estimated value {circumflex over (λ)}_(qdr) ^(s) estimated bythe current and flux estimation module 20 are inputted to the DFOCcalculation module 314.

The DFOC calculation module 314 internally generates a stator currentcommand value of they component and a stator current command value ofthe δ component based on the inputted values, and generates the voltagecommand value V_(γ)* of the γ component and the voltage command valueV_(δ)* of they component so that the γ-axis stator current i_(γ) and theδ-axis stator current i_(δ) follow the command values.

The voltage command value V_(γ)* of the γ component and the voltagecommand value V_(δ)* are inputted to the fifth coordinate conversionmodule 15 a.

The voltage command value V_(γ)* of they component and the voltagecommand value V_(δ)* re transmitted to the inverter 8 as gate pulses viathe PWM control module 16.

Since the torque command limit module 13 and the like are omitted, theadvantage of “operation at high power in a wider frequency range”, whichis achieved in the first embodiment, is restricted in the modificationof the third embodiment as compared with the first embodiment.

However, similarly to the second embodiment, the speed and phaseestimation module 21 is provided and an input value to the speed andphase estimation module 21 is the rotor dqs-axes flux estimated value{circumflex over (λ)}_(qdr) ^(S) in the third embodiment. Thus, theeffects of “improvement of calculation accuracy of the estimated speedvalue and sensor-less field weakening region increase by the use of therotor flux”, which are the same as those of the second embodiment, canbe obtained in the third embodiment.

It should be noted that, although the DB-DFTC calculation module 14 isreplaced with the DFOC calculation module 314 in the modification inFIG. 19, the DB-DFTC calculation module 14 may be replaced with a FOCcalculation module or an IFOC calculation module.

The controller 11 according to the third embodiment can be achieved byusing the structure illustrated in the hardware configuration diagram inFIG. 20.

The features and advantages of the present disclosure (or embodiments)may be summarized as follows.

In the first controller for a power convertor according to the firstaspect of the present disclosure, the upper limit torque value iscalculated to be smaller as the fundamental wave output frequency ofoutput from the power convertor increases in a field weakening region.In other words, the torque limiter range is set to be smaller as thefundamental wave output frequency increases. The range of the torquecommand value for maintaining control stability tends to be smaller in afaster operation region in which the fundamental wave output frequencyis larger. The torque limiter range is dynamically adjusted inaccordance with this tendency. The torque command value changes only inthe adjusted torque limiter range, and thus torque increase can beachieved without losing the control stability. As a result, the controlstability and the output torque increase can be simultaneously achieved.

According to the second controller for a power convertor according tothe second aspect of the present disclosure, the estimated value of therotor flux is used in calculation. The estimated value of the rotor fluxincludes less noise than the estimated value of the stator flux, andthus the estimated speed value can be highly accurately calculated withless ripple. The torque command value has less ripple.

According to the third aspect of the present disclosure, the motordriving system including the first controller for a power convertoraccording to the first aspect can simultaneously achieve the controlstability and the output torque increase when the motor is operated fastin the field weakening region. The motor driving system including thesecond controller for a power convertor according to the second aspectcan highly accurately calculate the estimated speed value.

The invention claimed is:
 1. A controller for a power convertor, thecontroller comprising: a torque command value calculation moduleconfigured to calculate a first torque command value to a powerconvertor based on a speed command value of a motor driven by the powerconvertor; a torque command limit module configured to receive the firsttorque command value and generate a second torque command value obtainedby correcting the first torque command value so that the first torquecommand value is limited to a torque limiter range defined by an upperlimit torque value calculated by a predetermined calculation formula forupper limit torque command value calculation and a lower limit torquevalue obtained by multiplying the upper limit torque value by apredetermined negative coefficient or zero; a flux command generationmodule configured to generate a stator flux command value in accordancewith a fundamental wave output frequency of output from the powerconvertor; and an output voltage command module configured to: calculatean output voltage command value of the power convertor based on thesecond torque command value and the stator flux command value; generatea pulse width modulation signal based on the output voltage commandvalue; and transmit the pulse width modulation signal, wherein thetorque command limit module calculates the upper limit torque value tobe smaller as the fundamental wave output frequency increases at leastin a speed region equal to or higher than a field weakening startingpoint.
 2. The controller for a power convertor according to claim 1,wherein the torque command limit module includes a first blockconfigured to calculate the upper limit torque value in accordance witha predetermined first calculation formula for upper limit torque commandvalue calculation when the fundamental wave output frequency belongs toa first speed region between the field weakening starting point and aboundary speed predetermined to be higher than the field weakeningstarting point, and the first calculation formula is a second-orderpolynomial including a second-order term and a first-order term of thestator flux command value.
 3. The controller for a power convertoraccording to claim 2, wherein the first block determines the torquelimiter range in a normal speed region lower than the field weakeningstarting point.
 4. The controller for a power convertor according toclaim 2, wherein the torque command limit module includes a second blockconfigured to calculate the upper limit torque value in accordance witha predetermined second calculation formula for upper limit torquecommand value calculation when the fundamental wave output frequencybelongs to a second speed region equal to or higher than the boundaryspeed predetermined to be higher than the field weakening startingpoint, and the second calculation formula is a monomial obtained bymultiplying the second-order term of the stator flux command value by apredetermined coefficient.
 5. The controller for a power convertoraccording to claim 1, wherein the flux command generation module limitsthe magnitude of the stator flux command value to be equal to or smallerthan a predetermined upper limiter flux value.
 6. A motor driving systemcomprising: a power convertor configured to drive a motor; and thecontroller according to claim 1 configured to control the powerconvertor.
 7. A controller for a power convertor, the controllercomprising: a torque command value calculation module configured tocalculate a torque command value to a power convertor based on a speedcommand value of a motor driven by the power convertor; a voltagecommand value calculation module configured to calculate a voltagecommand value to the power convertor based on the torque command valuecalculated by the torque command value calculation module; a fluxestimation module configured to calculate estimated values of the statorflux and rotor flux of the motor in a subsequent control period based onthe voltage command value to the power convertor and a measured statorcurrent of the motor; a motor speed estimation module configured tocalculate an estimated value of a speed of the motor in a subsequentcontrol period based on the estimated value of the rotor flux calculatedby the flux estimation module; and an inverter controller configured to:generate a pulse width modulation signal based on at least one of thetorque command value, the voltage command value, the estimated values ofthe stator flux, or the speed of the motor in a subsequent controlperiod; and transmit the pulse width modulation signal.
 8. Thecontroller for a power convertor according to claim 7, furthercomprising a switching module provided between the flux estimationmodule and the motor speed estimation module and configured toselectively transmit one of the estimated values of the stator flux androtor flux of the motor to the motor speed estimation module.
 9. A motordriving system comprising: a power convertor configured to drive amotor; and the controller according to claim 7 configured to control thepower convertor.
 10. The controller for a power convertor according toclaim 1, wherein: a first field weakening region and a second fieldweakening region are set; the upper limit torque value is calculatedaccording to a first calculation formula in the first field weakeningregion, and the upper limit limiter torque value is further calculatedaccording to a second calculation formula, different from the firstcalculation formula, in the second field weakening region.
 11. Thecontroller for a power convertor according to claim 1, wherein: a firstarithmetic expression calculates the upper limit torque value based onboth a current limit and a voltage limit of the power converter, and asecond arithmetic expression calculates the upper limit torque valuebased on only the voltage limit.
 12. The controller for a powerconvertor according to claim 11, wherein: the first arithmeticexpression differs from the second arithmetic expression based on thedifference between a first torque command value and a second torquecommand value; the first torque command value is defined as${{\frac{3}{2}\frac{P}{2}\frac{L_{m}^{2}}{\sigma\; L_{s}^{2}L_{r}}\lambda_{q{s\_ opt}}^{e}\lambda_{d{s\_ opt}}^{e}} = {\frac{3}{2}\frac{P}{2}\frac{L_{m}^{2}}{\left( {1 - {\sigma\;}^{2}} \right)L_{s}^{2}L_{r}}\sqrt{\left( {{L_{s}^{2}I_{smax}^{2}} - \lambda_{s\_ opt}^{2}} \right)\left( {\lambda_{s\_ opt}^{2} - {\sigma^{2}L_{s}^{2}I_{smax}^{2}}} \right)}}};$and the second torque command value is defined as$\frac{3}{4}\frac{P}{2}\frac{L_{m}^{2}}{\sigma\; L_{s}^{2}L_{r}}{\lambda_{s\_ opt}^{2}.}$13. The controller for a power convertor according to claim 10, wherein:the first field weakening region and the second field weakening regionare set so that in a graph having λ^(eds) and λ^(eqs) as axes, theintersection point between an ellipse, representing an inverter current,and a circle, representing and inverter voltage, is the second fieldweakening region in a region existing below the line of λ^(eds) andλ^(eqs).